ON Semiconductort Power Factor Controllers
The MC34262/MC33262 are active power factor controllers specifically designed for use as a preconverter in electronic ballast and in off­line power converter applications. These integrated circuits feature an internal startup timer for stand­alone applications, a one quadrant multiplier for near unity power factor, zero current detector to ensure critical conduction operation, transconductance error amplifier, quickstart circuit for enhanced startup, trimmed internal bandgap reference, current sensing comparator, and a totem pole output ideally suited for driving a power MOSFET. Also included are protective features consisting of an overvoltage comparator to eliminate runaway output voltage due to load removal, input undervoltage lockout with hysteresis, cycle­by­cycle current limiting, multiplier output clamp that limits maximum peak switch current, an RS latch for single pulse metering, and a drive output high state clamp for MOSFET gate protection. These devices are available in dual­in­line and surface mount plastic packages. · Overvoltage Comparator Eliminates Runaway Output Voltage · Internal Startup Timer · One Quadrant Multiplier · Zero Current Detector · Trimmed 2% Internal Bandgap Reference · Totem Pole Output with High State Clamp · Undervoltage Lockout with 6.0 V of Hysteresis · Low Startup and Operating Current · Supersedes Functionality of SG3561 and TDA4817
Simplified Block Diagram
Zero Current Detector 5 Zero Current Detect Input MC34262 MC33262
POWER FACTOR CONTROLLERS
SEMICONDUCTOR TECHNICAL DATA 8 1 P SUFFIX PLASTIC PACKAGE CASE 626 8 1 D SUFFIX PLASTIC PACKAGE CASE 751 (SO­8) PIN CONNECTIONS 2.5V Reference Undervoltage Lockout 8 VCC Voltage Feedback Input Compensation Multiplier Input Current Sense Input 1 2 3 4 (Top View) 8 VCC 7 Drive Output 6 Gnd 5 Zero Current Detect Input Multiplier, Latch, PWM, Timer, & Logic 7 4 Drive Output Current Sense Input Overvoltage Comparator + Error Amp 1.08 Vref + Vref ORDERING INFORMATION
Device MC34262D Operating Temperature Range TA = 0° to +85°C Package SO­8 Plastic DIP SO­8 Plastic DIP Multiplier Input 3 Multiplier Quickstart Gnd 6 Compensation 2 Voltage Feedback 1 Input MC34262P MC33262D MC33262P TA = ­40° to +105°C © Semiconductor Components Industries, LLC, 2001 1 March, 2001 ­ Rev. 2 Publication Order Number: MC34262/D MC34262 MC33262
MAXIMUM RATINGS
Rating Total Power Supply and Zener Current Output Current, Source or Sink (Note 1) Current Sense, Multiplier, and Voltage Feedback Inputs Zero Current Detect Input High State Forward Current Low State Reverse Current Power Dissipation and Thermal Characteristics P Suffix, Plastic Package, Case 626 Maximum Power Dissipation @ TA = 70°C Thermal Resistance, Junction­to­Air D Suffix, Plastic Package, Case 751 Maximum Power Dissipation @ TA = 70°C Thermal Resistance, Junction­to­Air Operating Junction Temperature Operating Ambient Temperature (Note 3) MC34262 MC33262 Storage Temperature Symbol (ICC + IZ) IO Vin Iin 50 ­10 Value 30 500 ­1.0 to +10 Unit mA mA V mA PD RJA PD RJA TJ TA 800 100 450 178 +150 0 to + 85 ­ 40 to +105 mW °C/W mW °C/W °C °C Tstg ­ 65 to +150 °C ELECTRICAL CHARACTERISTICS (VCC = 12 V (Note 2), for typical values TA = 25°C, for min/max values TA is the operating ambient temperature range that applies [Note 3], unless otherwise noted.)
Characteristic ERROR AMPLIFIER Voltage Feedback Input Threshold TA = 25°C TA = Tlow to Thigh (VCC = 12 V to 28 V) Line Regulation (VCC = 12 V to 28 V, TA = 25°C) Input Bias Current (VFB = 0 V) Transconductance (TA = 25°C) Output Current Source (VFB = 2.3 V) Sink (VFB = 2.7 V) Output Voltage Swing High State (VFB = 2.3 V) Low State (VFB = 2.7 V) OVERVOLTAGE COMPARATOR Voltage Feedback Input Threshold MULTIPLIER Input Bias Current, Pin 3 (VFB = 0 V) Input Threshold, Pin 2 Dynamic Input Voltage Range Multiplier Input (Pin 3) Compensation (Pin 2) Multiplier Gain (VPin 3 = 0.5 V, VPin 2 = Vth(M) + 1.0 V) (Note 4) ZERO CURRENT DETECTOR Input Threshold Voltage (Vin Increasing) Hysteresis (Vin Decreasing) Input Clamp Voltage High State (IDET = + 3.0 mA) Low State (IDET = ­ 3.0 mA) Vth VH VIH VIL 1.33 100 6.1 0.3 1.6 200 6.7 0.7 1.87 300 -- 1.0 V mV V IIB Vth(M) VPin 3 VPin 2 K -- 1.05 VOL(EA) 0 to 2.5 Vth(M) to (Vth(M) + 1.0) 0.43 ­ 0.1 1.2 VOL(EA) 0 to 3.5 Vth(M) to (Vth(M) + 1.5) 0.65 ­ 0.5 -- -- -- 0.87 1/V µA V V VFB(OV) 1.065 VFB 1.08 VFB 1.095 VFB V VFB 2.465 2.44 Regline IIB gm IO -- -- VOH(ea) VOL(ea) 5.8 -- 10 10 6.4 1.7 -- -- V -- 2.4 -- -- 80 2.5 -- 1.0 ­ 0.1 100 2.535 2.54 10 ­ 0.5 130 mV µA µmho µA V Symbol Min Typ Max Unit http://onsemi.com
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ELECTRICAL CHARACTERISTICS (VCC = 12 V (Note 2), for typical values TA = 25°C, for min/max values TA is the operating ambient temperature range that applies (Note 3), unless otherwise noted.)
Characteristic CURRENT SENSE COMPARATOR Input Bias Current (VPin 4 = 0 V) Input Offset Voltage (VPin 2 = 1.1 V, VPin 3 = 0 V) Maximum Current Sense Input Threshold (Note 5) Delay to Output DRIVE OUTPUT Output Voltage (VCC = 12 V) Low State (ISink = 20 mA) Low State (ISink = 200 mA) High State (ISource = 20 mA) High State (ISource = 200 mA) Output Voltage (VCC = 30 V) High State (ISource = 20 mA, CL = 15 pF) Output Voltage Rise Time (CL = 1.0 nF) Output Voltage Fall Time (CL = 1.0 nF) Output Voltage with UVLO Activated (VCC = 7.0 V, ISink = 1.0 mA) RESTART TIMER Restart Time Delay UNDERVOLTAGE LOCKOUT Startup Threshold (VCC Increasing) Minimum Operating Voltage After Turn­On (VCC Decreasing) Hysteresis TOTAL DEVICE Power Supply Current Startup (VCC = 7.0 V) Operating Dynamic Operating (50 kHz, CL = 1.0 nF) Power Supply Zener Voltage (ICC = 25 mA)
NOTES: 1. Maximum package power dissipation limits must be observed. 2. Adjust VCC above the startup threshold before setting to 12 V. 3. Tlow = 0°C for MC34262 3. Tlow = ­ 40°C for MC33262 Thigh = + 85°C for MC34262 Thigh = +105°C for MC33262 Symbol Min Typ Max Unit µA mV V ns IIB VIO Vth(max) tPHL(in/out) -- -- 1.3 -- ­ 0.15 9.0 1.5 200 ­1.0 25 1.8 400 V VOL VOH VO(max) 14 tr tf VO(UVLO) -- -- -- 16 50 50 0.1 18 120 120 0.5 ns ns V -- -- 9.8 7.8 0.3 2.4 10.3 8.4 0.8 3.3 -- -- V tDLY 200 620 -- µs Vth(on) VShutdown VH 11.5 7.0 3.8 13 8.0 5.0 14.5 9.0 6.2 V V V ICC -- -- -- VZ 30
4. K = mA 0.25 6.5 9.0 36
Pin 4 Threshold VPin 3 (VPin 2 ­ Vth(M)) 0.4 12 20 -- V 5. This parameter is measured with V FB = 0 V, and VPin 3 = 3.0 V Figure 1. Current Sense Input Threshold versus Multiplier Input
V CS , CURRENT SENSE PIN 4 THRESHOLD (V) V CS , CURRENT SENSE PIN 4 THRESHOLD (V) 1.6 1.4 1.2 1.0 VCC = 12 V TA = 25°C VPin 2 = 3.75 V VPin 2 = 3.5 V VPin 2 = 2.75 V VPin 2 = 2.5 V VPin 2 = 2.25 V 0.08 0.07 Figure 2. Current Sense Input Threshold versus Multiplier Input, Expanded View
VPin 2 = 3.75 V VPin 2 = 3.5 V VPin 2 = 3.25 V 0.06 VPin 2 = 3.0 V 0.05 VPin 2 = 2.75 V 0.04 0.03 0.02 0.01 0 - 0.12 VPin 2 = 2.0 V - 0.06 0 0.06 0.12 0.18 VM, MULTIPLIER PIN 3 INPUT VOLTAGE (V) 0.24 VCC = 12 V TA = 25°C VPin 2 = 2.5 V VPin 2 = 2.25 V VPin 2 = 3.25 V 0.8 VPin 2 = 3.0 V 0.6 0.4 0.2 0 - 0.2 VPin 2 = 2.0 V 0.6 1.4 2.2 3.0 VM, MULTIPLIER PIN 3 INPUT VOLTAGE (V) 3.8 http://onsemi.com
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V FB , VOLTAGE FEEDBACK THRESHOLD CHANGE (mV) V FB(OV) , OVERVOLTAGE INPUT THRESHOLD (%VFB ) Figure 3. Voltage Feedback Input Threshold Change versus Temperature
4.0 0 VCC = 12 V Pins 1 to 2 Figure 4. Overvoltage Comparator Input Threshold versus Temperature
110 VCC = 12 V 109 - 4.0 - 8.0 -12 -16 - 55 108 107 - 25 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 100 125 106 - 55 - 25 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 100 125 Figure 5. Error Amp Transconductance and Phase versus Frequency
120 g , TRANSCONDUCTANCE (µ mho) m 100 80 60 40 20 0 3.0 k 10 k 30 k 100 k 300 k f, FREQUENCY (Hz) 1.0 M O , EXCESS PHASE (DEGREES) Phase Transconductance VCC = 12 V VO = 2.5 V to 3.5 V RL = 100 k to 3.0 V CL = 2.0 pF TA = 25°C 0 30 60 90 4.00 V Figure 6. Error Amp Transient Response
VCC = 12 V RL = 100 k CL = 2.0 pF TA = 25°C 0.75 V/DIV 5.0 µs/DIV 3.25 V 120 150 180 3.0 M 2.50 V Figure 7. Quickstart Charge Current versus Temperature
Vchg , QUICKSTART CHARGE VOLTAGE (V) 1.80 t DLY, RESTART TIME DELAY (µ s) VCC = 12 V 900 I chg , QUICKSTART CHARGE CURRENT (µ A) 800 Figure 8. Restart Timer Delay versus Temperature
VCC = 12 V 1.76 800 700 1.72 Voltage Current 700 600 1.68 600 500 1.64 - 55 - 25 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 100 500 125 400 - 55 - 25 0 25 50 75 100 125 TA, AMBIENT TEMPERATURE (°C) http://onsemi.com
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Figure 9. Zero Current Detector Input Threshold Voltage versus Temperature
Vsat , OUTPUT SATURATION VOLTAGE (V) 1.7 V th , THRESHOLD VOLTAGE (V) Upper Threshold (Vin, Increasing) VCC = 12 V 0 - 2.0 - 4.0 - 6.0 4.0 2.0 0 0 Sink Saturation (Load to VCC) Gnd 80 160 240 IO, OUTPUT LOAD CURRENT (mA) 320 Source Saturation (Load to Ground) Figure 10. Output Saturation Voltage versus Load Current
VCC VCC = 12 V 80 µs Pulsed Load 120 Hz Rate 1.6 1.5 1.4 Lower Threshold (Vin, Decreasing) - 25 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 100 125 1.3 - 55 Figure 11. Drive Output Waveform
90% VCC = 12 V CL = 1.0 nF TA = 25°C VO , OUTPUT VOLTAGE Figure 12. Drive Output Cross Conduction
5.0 V/DIV 125 100 mA/DIV 100 ns/DIV VCC = 12 V CL = 15 pF TA = 25°C 10% 100 ns/DIV Figure 13. Supply Current versus Supply Voltage
16 I CC , SUPPLY CURRENT (mA) VCC , SUPPLY VOLTAGE (V) 14 13 12 11 10 9.0 8.0 40 7.0 - 55 I CC , SUPPLY CURRENT Figure 14. Undervoltage Lockout Thresholds versus Temperature 12 Startup Threshold (VCC Increasing) 8.0 VFB = 0 V Current Sense = 0 V Multiplier = 0 V CL = 1.0 nF f = 50 kHz TA = 25°C 0 10 20 30 VCC, SUPPLY VOLTAGE (V) 4.0 Minimum Operating Threshold (VCC Decreasing) 0 - 25 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 100 http://onsemi.com
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FUNCTIONAL DESCRIPTION
Introduction With the goal of exceeding the requirements of legislation on line­current harmonic content, there is an ever increasing demand for an economical method of obtaining a unity power factor. This data sheet describes a monolithic control IC that was specifically designed for power factor control with minimal external components. It offers the designer a simple, cost­effective solution to obtain the benefits of active power factor correction. Most electronic ballasts and switching power supplies use a bridge rectifier and a bulk storage capacitor to derive raw dc voltage from the utility ac line, Figure 15.
Figure 15. Uncorrected Power Factor Circuit
Rectifiers AC Line + Converter appear resistive to the ac line, thus significantly reducing the harmonic current content.
Figure 16. Uncorrected Power Factor Input Waveforms
Vpk Rectified DC 0 Line Sag AC Line Voltage 0 Bulk Storage Capacitor Load AC Line Current This simple rectifying circuit draws power from the line when the instantaneous ac voltage exceeds the capacitor voltage. This occurs near the line voltage peak and results in a high charge current spike, Figure 16. Since power is only taken near the line voltage peaks, the resulting spikes of current are extremely nonsinusoidal with a high content of harmonics. This results in a poor power factor condition where the apparent input power is much higher than the real power. Power factor ratios of 0.5 to 0.7 are common. Power factor correction can be achieved with the use of either a passive or an active input circuit. Passive circuits usually contain a combination of large capacitors, inductors, and rectifiers that operate at the ac line frequency. Active circuits incorporate some form of a high frequency switching converter for the power processing, with the boost converter being the most popular topology, Figure 17. Since active input circuits operate at a frequency much higher than that of the ac line, they are smaller, lighter in weight, and more efficient than a passive circuit that yields similar results. With proper control of the preconverter, almost any complex load can be made to The MC34262, MC33262 are high performance, critical conduction, current­mode power factor controllers specifically designed for use in off­line active preconverters. These devices provide the necessary features required to significantly enhance poor power factor loads by keeping the ac line current sinusoidal and in phase with the line voltage.
Operating Description The MC34262, MC33262 contain many of the building blocks and protection features that are employed in modern high performance current mode power supply controllers. There are, however, two areas where there is a major difference when compared to popular devices such as the UC3842 series. Referring to the block diagram in Figure 19, note that a multiplier has been added to the current sense loop and that this device does not contain an oscillator. The reasons for these differences will become apparent in the following discussion. A description of each of the functional blocks is given below. Figure 17. Active Power Factor Correction Preconverter
Rectifiers AC Line + High Frequency Bypass Capacitor PFC Preconverter Converter + MC34362 Bulk Storage Capacitor Load http://onsemi.com
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The ac full wave rectified haversines are monitored at Pin 3 with respect to ground while the Error Amp output at Pin 2 is monitored with respect to the Voltage Feedback Input threshold. The Multiplier is designed to have an extremely linear transfer curve over a wide dynamic range, 0 V to 3.2 V for Pin 3, and 2.0 V to 3.75 V for Pin 2, Figure 1. The Multiplier output controls the Current Sense Comparator threshold as the ac voltage traverses sinusoidally from zero to peak line, Figure 18. This has the effect of forcing the MOSFET on­time to track the input line voltage, resulting in a fixed Drive Output on­time, thus making the preconverter load appear to be resistive to the ac line. An approximation of the Current Sense Comparator threshold can be calculated from the following equation. This equation is accurate only under the given test condition stated in the electrical table.
VCS, Pin 4 Threshold 0.65 (VPin 2 ­ Vth(M)) VPin 3 Error Amplifier An Error Amplifier with access to the inverting input and output is provided. The amplifier is a transconductance type, meaning that it has high output impedance with controlled voltage­to­current gain. The amplifier features a typical gm of 100 µmhos (Figure 5). The noninverting input is internally biased at 2.5 V ± 2.0% and is not pinned out. The output voltage of the power factor converter is typically divided down and monitored by the inverting input. The maximum input bias current is ­ 0.5 µA, which can cause an output voltage error that is equal to the product of the input bias current and the value of the upper divider resistor R2. The Error Amp output is internally connected to the Multiplier and is pinned out (Pin 2) for external loop compensation. Typically, the bandwidth is set below 20 Hz, so that the amplifier's output voltage is relatively constant over a given ac line cycle. In effect, the error amp monitors the average output voltage of the converter over several line cycles. The Error Amp output stage was designed to have a relatively constant transconductance over temperature. This allows the designer to define the compensated bandwidth over the intended operating temperature range. The output stage can sink and source 10 µA of current and is capable of swinging from 1.7 V to 6.4 V, assuring that the Multiplier can be driven over its entire dynamic range. A key feature to using a transconductance type amplifier, is that the input is allowed to move independently with respect to the output, since the compensation capacitor is connected to ground. This allows dual usage of of the Voltage Feedback Input pin by the Error Amplifier and by the Overvoltage Comparator.
Overvoltage Comparator A significant reduction in line current distortion can be attained by forcing the preconverter to switch as the ac line voltage crosses through zero. The forced switching is achieved by adding a controlled amount of offset to the Multiplier and Current Sense Comparator circuits. The equation shown below accounts for the built­in offsets and is accurate to within ten percent. Let Vth(M) = 1.991 V
VCS, Pin 4 Threshold = 0.544 (VPin 2 ­ Vth(M)) VPin 3 + 0.0417 (VPin 2 ­ Vth(M)) Zero Current Detector An Overvoltage Comparator is incorporated to eliminate the possibility of runaway output voltage. This condition can occur during initial startup, sudden load removal, or during output arcing and is the result of the low bandwidth that must be used in the Error Amplifier control loop. The Overvoltage Comparator monitors the peak output voltage of the converter, and when exceeded, immediately terminates MOSFET switching. The comparator threshold is internally set to 1.08 Vref. In order to prevent false tripping during normal operation, the value of the output filter capacitor C3 must be large enough to keep the peak­to­peak ripple less than 16% of the average dc output. The Overvoltage Comparator input to Drive Output turn­off propagation delay is typically 400 ns. A comparison of startup overshoot without and with the Overvoltage Comparator circuit is shown in Figure 23.
Multiplier A single quadrant, two input multiplier is the critical element that enables this device to control power factor. The MC34262 operates as a critical conduction current mode controller, whereby output switch conduction is initiated by the Zero Current Detector and terminated when the peak inductor current reaches the threshold level established by the Multiplier output. The Zero Current Detector initiates the next on­time by setting the RS Latch at the instant the inductor current reaches zero. This critical conduction mode of operation has two significant benefits. First, since the MOSFET cannot turn­on until the inductor current reaches zero, the output rectifier reverse recovery time becomes less critical, allowing the use of an inexpensive rectifier. Second, since there are no deadtime gaps between cycles, the ac line current is continuous, thus limiting the peak switch to twice the average input current. The Zero Current Detector indirectly senses the inductor current by monitoring when the auxiliary winding voltage falls below 1.4 V. To prevent false tripping, 200 mV of hysteresis is provided. Figure 9 shows that the thresholds are well­defined over temperature. The Zero Current Detector input is internally protected by two clamps. The upper 6.7 V clamp prevents input overvoltage breakdown while the lower 0.7 V clamp prevents substrate injection. Current limit protection of the lower clamp transistor is provided in the event that the input pin is accidentally shorted to ground. The Zero Current Detector input to Drive Output turn­on propagation delay is typically 320 ns. http://onsemi.com
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Figure 18. Inductor Current and MOSFET Gate Voltage Waveforms
Peak automatically start or restart the preconverter if the Drive Output has been off for more than 620 µs after the inductor current reaches zero. The restart time delay versus temperature is shown in Figure 8.
Undervoltage Lockout and Quickstart Inductor Current Average 0 On MOSFET Q1 Off Current Sense Comparator and RS Latch The Current Sense Comparator RS Latch configuration used ensures that only a single pulse appears at the Drive Output during a given cycle. The inductor current is converted to a voltage by inserting a ground­referenced sense resistor R7 in series with the source of output switch Q1. This voltage is monitored by the Current Sense Input and compared to a level derived from the Multiplier output. The peak inductor current under normal operating conditions is controlled by the threshold voltage of Pin 4 where:
IL(pk ) = Pin 4 Threshold R7 Abnormal operating conditions occur during preconverter startup at extremely high line or if output voltage sensing is lost. Under these conditions, the Multiplier output and Current Sense threshold will be internally clamped to 1.5 V. Therefore, the maximum peak switch current is limited to:
Ipk(max) = 1.5 V R7 An Undervoltage Lockout comparator has been incorporated to guarantee that the IC is fully functional before enabling the output stage. The positive power supply terminal (VCC) is monitored by the UVLO comparator with the upper threshold set at 13 V and the lower threshold at 8.0 V. In the stand­by mode, with VCC at 7.0 V, the required supply current is less than 0.4 mA. This large hysteresis and low startup current allow the implementation of efficient bootstrap startup techniques, making these devices ideally suited for wide input range off­line preconverter applications. An internal 36 V clamp has been added from VCC to ground to protect the IC and capacitor C4 from an overvoltage condition. This feature is desirable if external circuitry is used to delay the startup of the preconverter. The supply current, startup, and operating voltage characteristics are shown in Figures 13 and 14. A Quickstart circuit has been incorporated to optimize converter startup. During initial startup, compensation capacitor C1 will be discharged, holding the error amp output below the Multiplier threshold. This will prevent Drive Output switching and delay bootstrapping of capacitor C4 by diode D6. If Pin 2 does not reach the multiplier threshold before C4 discharges below the lower UVLO threshold, the converter will "hiccup" and experience a significant startup delay. The Quickstart circuit is designed to precharge C1 to 1.7 V, Figure 7. This level is slightly below the Pin 2 Multiplier threshold, allowing immediate Drive Output switching and bootstrap operation when C4 crosses the upper UVLO threshold.
Drive Output An internal RC filter has been included to attenuate any high frequency noise that may be present on the current waveform. This filter helps reduce the ac line current distortion especially near the zero crossings. With the component values shown in Figure 20, the Current Sense Comparator threshold, at the peak of the haversine varies from 1.1 V at 90 Vac to 100 mV at 268 Vac. The Current Sense Input to Drive Output turn­off propagation delay is typically less than 200 ns.
Timer A watchdog timer function was added to the IC to eliminate the need for an external oscillator when used in stand­alone applications. The Timer provides a means to The MC34262/MC33262 contain a single totem­pole output stage specifically designed for direct drive of power MOSFETs. The Drive Output is capable of up to ± 500 mA peak current with a typical rise and fall time of 50 ns with a 1.0 nF load. Additional internal circuitry has been added to keep the Drive Output in a sinking mode whenever the Undervoltage Lockout is active. This characteristic eliminates the need for an external gate pull­down resistor. The totem­pole output has been optimized to minimize cross­conduction current during high speed operation. The addition of two 10 resistors, one in series with the source output transistor and one in series with the sink output transistor, helps to reduce the cross­conduction current and radiated noise by limiting the output rise and fall time. A 16 V clamp has been incorporated into the output stage to limit the high state VOH. This prevents rupture of the MOSFET gate when VCC exceeds 20 V. http://onsemi.com
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APPLICATIONS INFORMATION The application circuits shown in Figures 19, 20 and 21 reveal that few external components are required for a complete power factor preconverter. Each circuit is a peak detecting current­mode boost converter that operates in critical conduction mode with a fixed on­time and variable off­time. A major benefit of critical conduction operation is that the current loop is inherently stable, thus eliminating the need for ramp compensation. The application in Figure 19 operates over an input voltage range of 90 Vac to 138 Vac and provides an output power of 80 W (230 V at 350 mA) with an associated power factor of approximately
Notes
Calculate the maximum required output power. Calculated at the minimum required ac line voltage for output regulation. Let the efficiency = 0.92 for low line operation. Let the switching cycle t = 40 µs for universal input (85 to 265 Vac) operation and 20 µs for fixed input (92 to 138 Vac, or 184 to 276 Vac) operation. In theory the on­time ton is constant. In practice ton tends to increase at the ac line zero crossings due to the charge on capacitor C5. Let Vac = Vac(LL) for initial ton and toff calculations. The off­time toff is greatest at the peak of the ac line voltage and approaches zero at the ac line zero crossings. Theta () represents the angle of the ac line voltage. The minimum switching frequency occurs at the peak of the ac line voltage. As the ac line voltage traverses from peak to zero, toff approaches zero producing an increase in switching frequency. Set the current sense threshold VCS to 1.0 V for universal input (85 Vac to 265 Vac) operation and to 0.5 V for fixed input (92 Vac to 138 Vac, or 184 Vac to 276 Vac) operation. Note that VCS must be