ction converter with EMI filter
VOL. 2, NO. 4, AUGUST 2007 ISSN 1819-6608
ARPN Journal of Engineering and Applied Sciences

©2006-2007 Asian Research Publishing Network (ARPN). All rights reserved.

www.arpnjournals.com


SIMULATION OF A NOVEL ZERO VOLTAGE TRANSITION TECHNIQUE
BASED ON BOOST POWER FACTOR CORRECTION CONVERTER
WITH EMI FILTER

P. Ram Mohan
1
, M. Vijaya Kumar
2
, O.V. Raghava Reddy
3
and S. Rama Reddy
4
1
G. Pulla Reddy Engineering College, Kurnool, Andhra Pradesh, India
2
JNTU College of Engineering, Anantapur, Andhra Pradesh, India
3
ISRO Satellite Center, Bangalore, Karnataka, India
4
Jerusalem College of Engineering, Chennai, Tamil Nadu, India
E-mail:
rammohan_cdp@yahoo.co.in


ABSTRACT
A novel Zero Voltage Transition (ZVT) technique based closed loop control of Boost Power Factor Correction
(PFC) converter with Electro Magnetic Interference (EMI) Filter is presented in this paper. It
operates at a fixed frequency
while achieving zero voltage turn-on of the main switch and zero current turn-off of the boost diode. This is accomplished
by employing resonant operation only during switch transitions. During the rest of the cycle, the resonant network is
essentially removed from the circuit and converter operation is identical to its non-resonant counterpart.
The principle of
operation and simulation results of proposed converter is presented in this paper. The power factor is improved to near 0.99
using the proposed converter.

Keywords: z
ero voltage transition, power factor correction, electro magnetic interference, common mode, differential mode.

INTRODUCTION
The demand for power, which is increased
tremendously over the last few decades, has forced the
power engineers to establish reliable network in order to
supply quality power to the consumers. Power factor,
which is defined as the cosine of the phase angle between
the voltage and current signals, plays a key role in
delivering quality power to the consumers [1].
Over the years lot of research has been carried
out for the control of the power factor. This research got a
tremendous
boost with the strides made in the
miniaturization of the electronic industry. The component
of input current normal to voltage across the load
resistance wastes power in the resistance of the source
generator. In power supplies with a capacitor filter across
the input bridge rectifier, the input line current consists of
very narrow spikes with the fast rise and fall time. These
current spikes have a high rms value, waste power and
give rise to RFI/EMI problems. Power supplies with such
input line currents have poor power factor. Power Factor
Correction seeks to eliminate such line current spikes and
force input current to be sinusoidal, in phase with input
voltage and to generate a fairly well regulated DC output
voltage somewhat greater than the peak of the incoming
sine wave [2,3].
Generally EMI problems arise due to the sudden
changes in voltage (dv/dt) or current (di/dt) levels in a
waveform. In diode rectifier, the line current can be pulse
of short duration and the diode recovery current pulse can
generate transient voltage spikes in the line inductance. A
conductor carrying dv/dt wave acts like an antenna and
sensitive signal circuit and appear as noise. The EMI
problems create communication line interference with
sensitive signal electronic circuits.



Basic boost power factor correction converter
Boost converters can be operated in either the
discontinuous or continuous mode. But the continuous-
mode boost topology is far better suited to yield relatively
smooth, ripple-free half sinusoids of input line current in
this application. This can be seen from Figure 1, which
shows a continuous-mode boost converter fed from a
constant DC input voltage. The continuous-mode boost
topology differs significantly from the discontinuous
mode.
In the discontinuous mode, the inductor L
1
is
made small to yield a steep ramp (di/dt = V
IN
/L
1
) of input
current to Q
1
. When Q
1
turns off, all the current or energy
stored in L
1
is transferred via D
1
to the load. Since L
1
is
small, the downward ramp of current through D
1
[di/dt =
(V
0
-V
IN
)/L
1
] is also steep and D
1
current falls to zero
before the next Q
1
turn-on. The input line current then,
which is the sum of the Q
1
current when it is ON and the
D
1
current when Q
1
is OFF, is not at all constant over one
complete switching cycle. It consists of steep up and down
ramps with zero current gaps between the end of a turnoff
and the next turn-on.
But in the continuous-mode of Figure-1, the
inductor L
1
is made significantly larger. Then the Q
1

current (Figure-1c) has the shape of a large step of current
with a slow upward ramp on it, and the D
1
current has the
shape of a large step with a slow downward ramp. And
importantly, there is no gap of zero current between the
end of a turnoff and the next turn-on. Input line current
(Figure-1e) is now the sum of the I
Q1
and I
D1
currents and
if the ramps are made small by using a large L
1
, the line
input current over one switching cycle is then a constant
I
av
with very small peak-to-peak ripple of I. The input
power is V
IN
I
av
.


1
VOL. 2, NO. 4, AUGUST 2007 ISSN 1819-6608
ARPN Journal of Engineering and Applied Sciences

©2006-2007 Asian Research Publishing Network (ARPN). All rights reserved.

www.arpnjournals.com


Now for an Ac input, such a continuous-mode
boost converter is used after the input bridge rectifier
output. At any point on the half sinusoid input voltage, the
Q
1
ON time will be forced by a PWM control chip to
boost that instantaneous voltage to the desired DC output
voltage. A DC voltage error amplifier, a DC reference
voltage, and a pulse with modulator in the control chip
modulate the Q
1
ON time in a negative feedback loop, to
yield a constant DC output voltage throughout the half
sinusoidal input voltage. The instantaneous input line
current will be sensed by a sensing resistor R
S
and will be
proportional to the instantaneous voltage throughout the
half sinusoid. During any one ON time, current flows
through L
1
, Q
1
and R
S
back to the negative end of the
bridge and during the following OFF time it flows through
L
1
, D
1
, R
0
and C
0
in parallel and R
S
back to the negative
end of the bridge. By making L
1
large, the peak-to-peak
ripple current throughout each switching cycle is kept
small. Depending on switching speed of Q
1
, there may be
very narrow spikes on the half sinusoids of current
monitored in R
S
. If present, these may cause an RFI
problem. But a very small capacitor (in the vicinity of
1.0mF) across R
S
easily eliminates them.


L
1
D
1
C
0
0
R
Q
1
+
-
EA
V
ref
+
-
PWM
V
TPB
V
PWMO
V
b
eao
V
ea in
V
IN
V
b
V
0
T
ON
V
eao
V
t
V
PWMO

Figure-1a


V (Q1)
b

Figure-1b


I
av
I(Q1)

Figure-1c


I(D1)

Figure-1d

I
IN
amperes
0

Figure-1e

I(Q1)
for various DC load
currents at a fixed input
voltage

Figure-1f
Figure-1. Continuous conduction mode Boost Converter
and wave forms of Q
1
and D
1
for various DC load currents
at a fixed DC input voltage.

Zero Voltage Transition (ZVT) technique
The power stage of proposed PFC converter with
ZVT technique based boost topology is shown in Figure-2
and the ZVT timing diagram is shown in Figure-3. The
capacitive turn-on losses can be theoretically eliminated
and the overlap of non-negligible active switch voltage
and current can be avoided at turn-on, by using the Zero
Voltage Switching ZVS technique. Basically, this
technique consists of forcing to zero the active switch
voltage, prior to its turn-on, by creating a resonance
between an inductor and a capacitor. The inductor also
limits the rate of variation of the diode current, so losses
due to the reverse recovery are reduced as well [4].
However, better characteristics are obtained in
Zero Voltage Transition ZVT topologies, at the expense
of increased complexity. Here, to achieve ZVS, switch
voltage and current waveforms are changed only during
commutation intervals, the behavior of the ZVT converter
being otherwise identical to that of the hard-switching
converter.
In converter topologies having only one active
switch, the ZVT technique is implemented with an
auxiliary circuit, which consists of an additional active
switch, an auxiliary inductor, for the resonant process that
discharges the drain-source capacitance of the switch and
for limiting the rate of change of the diode current at turn-
off, as well as a few other passive components [5,6].
The auxiliary switch is turned on before turning
on the main active switch. This initiates a resonant
process, which creates zero voltage switching conditions
for the main active switch. The time intervals where the
2
VOL. 2, NO. 4, AUGUST 2007 ISSN 1819-6608
ARPN Journal of Engineering and Applied Sciences

©2006-2007 Asian Research Publishing Network (ARPN). All rights reserved.

www.arpnjournals.com