Optimizing the Ultra-Fast POWERplanar Rectifier Diode for Switching Power Supplies Optimizing the Ultra-Fast POWERplanar TM Rectifier Diode for Switching Power Supplies
INTRODUCTION A key device in all high voltage AC-DC power supplies is the ultrafast reverse recovery rectifier diode These diodes (D1 and D2 in Figure 1 ) not only play a major role in power supply efficiency but also can be major contributors to circuit electromagnetic interference (EMI) and even cause transistor failure if they are not selected correctly One would assume that by now this rectifier diode should approximate the behavior of an ideal switch i e zero on-state voltage no reverse leakage current and instanteous turnon At first glance the design of this single pn-junction device would appear to be quite straight forward but a review of the device equations reveals that many compromises must be made to optimize its performance An understanding of these tradeoffs will allow the circuit designer to select the most appropriate rectifier diode National Semiconductor Application Note 557 Ralph E Locher January 1989 Consider how the non-ideal behavior of rectifier D2 affects the circuit performance of the buck regulator in Figure 1a The solid lines in Figure 2a depict the switching behavior of the transistor switch and rectifier in comparison to the waveforms (dashed lines) that represent an ideal rectifier There are four differences between the two cases 1 The most significant difference is that the peak collector current of the transistor switch (IT in Figure 2a ) at the end of turn-on (time t2) has been increased by the magnitude of the peak reverse recovery current of the rectifier (IR(REC)) Correspondingly the peak power dissipation within the transistor has increased from PT to PT as shown in Figure 2c 2 The maximum transistor voltage VT at turn-off (t4 ­t6 in Figure 2a ) has been increased by the dynamic voltage drop of the rectifier during turn-on Since buck regulators generally run at low voltages this increase has a minimal effect However it is more significant in the forward converter circuit of Figure 1b and in bridge circuits operating from high bus voltages where the voltage margins cannot be as generous 3 Since the rectifier is not ideal its power dissipation consists of the following components TL G 10062 ­ 1 a Conduction loss (VF x IF) during the on-time b Turn-off loss during time t2 ­t3 and turn-on loss during time t5 ­t6 (Figure 2d) c Reverse blocking loss (VR x IR) during period t3 ­t5 4 The rectifier regains its reverse blocking capability at time t2 A ``snappy'' rectifier that quickly turns off IR(REC) will contribute much more EMI than a ``soft'' fast recovery rectifier A better transistor switch will intensify rather than improve the shortcomings of the fast recovery rectifier so it is necessary to consider more fully the conduction and switching behavior of the rectifier diode FIGURE 1a Buck Regulator to Step-Down Input Voltage VIN TL G 10062 ­ 2 FIGURE 1b Forward Converter AN-557 POWERplanarTM is a trademark of National Semiconductor Corporation C1995 National Semiconductor Corporation TL G 10062 RRD-B30M115 Printed in U S A CONDUCTION LOSSES DC conduction or on-state losses occur whenever the rectifier is conducting forward current and consists simply of the integration of IF x VF during the on-time Literature has dealt extensively with the computation of VF for many different rectifier structures (Reference 1) The National Semiconductor POWERplanarTM line of fast recovery diodes are planar passivated P a N b N a epitaxial type for which a cross-sectional view can be found in Figure 4 It can be shown that VF is inversely proportional to minority carrier lifetime and directly proportional to epitaxial thickness (Wi in Figure 4 ) Figure 5 plots theoretical curves of normalized VF vs minority carrier lifetimes for rectifiers with 250V and 500V avalanche voltage breakdown Since trr is approximately equal to minority carrier lifetime it is apparent that high current pn-junction rectifiers are limited to 20 ns ­ 50 ns reverse recovery times because VF dramatically increases for minority carrier lifetimes less than these It is also apparent that the VF curves have a broad minima around 10 ns ­ 30 ns so that another reason to select this value of minority carrier lifetime is that VF becomes independent of small changes in minority carrier lifetime due to manufacturing tolerances It is immediately obvious that the key to maximizing current through the rectifier is to minimize VF However at 200 kHz reverse recovery losses will quadruple to 4W so that increasing attention must be paid to this parameter as operating frequency is raised TL G 10062­3 FIGURE 2 Transistor and Rectifier Voltage and Current Waveforms for the Buck Regulator in Figure 1a a) Transistor and Rectifier Voltage Waveforms b) Transistor and Rectifier Current Waveforms c) Transistor Power Dissipation d) Rectifier Power Dissipation POWER LOSSES IN THE ULTRA-FAST RECTIFIER DIODE Consider the idealized rectifier current and voltage waveforms in Figure 3 for a 50 kHz buck regulator Power dissipation within the rectifier for a 50% duty factor is P e P(conduction) a P(blocking) a P(reverse recovery) 1 P e (VFIF a VRIR) a VRMIR(REC)tbf 2 Typical values for a 200V 8A rectifier are IR e 1 mA f e 50 kHz VF e 0 9V tB e 25 ns (assuming tb e trr 2) IF e 8A VR e 50V IR(REC) e 2 0A VRM e 200V Pe 1 (8A) (0 9V) a (50V) (1 mA) 2 a (200V) (2A) (25 ns) (50 kHz) TL G 10062 ­ 4 FIGURE 3 Representative Current and Voltage Waveforms for the Rectifier in the Buck Regulator Found in Figure 1a P e 3 6W a 0 025W a 0 5W e 4 125W 2 TL G 10062 ­ 5 FIGURE 4 Cross-Sectional View of a POWERplanarTM P a N b N a Fast Recovery Rectifier REVERSE RECOVERY LOSSES All pn-junction rectifiers operating in the forward direction store charge in the form of excess minority carriers The amount of stored charge is proportional to the magnitude of the forward current The process by which a rectifier diode is brought out of conduction and returned to its block state is called commutation Figure 7 shows an expanded view of current commutation also called reverse recovery Starting at time t0 the rectifier is switched from its forward conducting state at a specified current ramp rate (bdIF dt) The current ramp rate will be determined by the external circuit (E L) or the turn-on time of a transistor switch During the time t1 ­t2 the store charge within the rectifier is able to supply more current than the circuit requires so that the rectifier behaves like a short circuit Stored charge is depleted both by the reverse recovery current and recombination within the rectifier Eventually the stored charge dwindles to the point that a depletion region around the junction starts to grow allowing the rectifier to regain its reverse blocking voltage capability (t2) From a circuit-design standpoint the most important parameters are the peak reverse recovery current and ``S'' the softness factor A ``snappy'' rectifier will produce a large amplitude voltage transient and contribute significantly to electro-magnetic interference Figure 8 illustrates the actual reverse recovery of two rectifier diodes The peak voltage of the snappy rectifier is 175V compared to 142V peak for the FRP820 the higher voltage resulting from both the higher IR(REC) and the fact that the reverse recovery current decays to zero in a shorter time TL G 10062 ­ 6 FIGURE 5 Normalized VF for 250V and 500V Rated Rectifiers as a Function of Minority Carrier Lifetime REVERSE BLOCKING LOSSES Planar passivation techniques have reduced surface leakage currents (IR) to a negligible amount so that the principle reverse leakage current is recombination current in the space charge region Some of the many methods to control minority carrier lifetimes are electron or neutron irradiation and gold or platinum diffusion each with its own advantages and disadvantages For 200V ultrafast recovery rectifiers gold diffusion still represents the best compromise between speed VF IR and ``soft'' recovery A drawback to gold diffusion is its relatively high reverse leakage current It should be pointed out that the reliability of the gold-diffused product is the same as other rectifiers (all other factors being equal) since this leakage current is a bulk and not a surface phenomenon Figure 6 illustrates the dependency of recombination current on junction temperature and minority carrier lifetime which is inversely proportional to the amount of gold in the depletion region Experimental leakage test results have been plotted in Figure 6 for the National Semiconductor 8A and 16A series of rectifiers (FRP820 and FRP1620 respectively) at 100 C 125 C and 150 C These points indicate that the low current injection level lifetime ranges from 20 ns­30 ns and is relatively independent of TJ Since reliability design guidelines specify that the rectifiers be operated at one-half their voltage rating and 25 C ­ 50 C below their maximum junction temperature the expected leakage currents in well designed power supplies will run less than 1 mA TL G 10062 ­ 7 FIGURE 6 Regeneration Current for Gold-Doped P a N b N a Rectifier Diodes 3 TL G 10062 ­ 8 FIGURE 7 Expanded View of Current Commutation in a Rectifier Diode The relative snappiness of a rectifier may be defined quantitatively by dividing the reverse recovery time trr into two subperiods ta and tb as shown in Figure 7 The softness factor ``S'' is simply the ratio tb ta A rectifier with a low value S factor will be more likely to produce dangerous voltage transients but it will also dissipate less reverse recovery energy than a high S factor rectifier A reasonable compromise between these two conflicting constraints would be to design a rectifier with S e 1 (ta e tb) The S factors of the FRP820 rectifier and the competitive device in Figure 8 are 0 55 and 0 31 respectively Only recently has it become possible to model the ramp recovery in p-i-n rectifiers (References 2 3) and the following equations have proved useful in predicting reverse recovery parameters trr e Se Wi0u Da 8 Wi 40Dau TL G 10062 ­ 9 Test Conditions Tj e 25 C IF e 8A dIF dt e 100 A mS I e 0 5 A DIV T e 10 ns DIV TL G 10062 ­ 10 Test Conditions IR(REC) e QR(REC) e where J dI 05u dt J
dIF u dt
2 F Wi 1a 80Dau J b1 Tj e 25 C IF e 8A dIF dt e 100 A ms I e 50 VA DIV T e 10 ns DIV FIGURE 8 Comparison of Reverse Recovery of the FRP820 Series Rectifier to a Snappy Rectifier REVERSE RECOVERY CHARACTERIZATION Figures 10­13 plot QR(REC) IR(REC) trr and S versus dIF dt for the FRP1600 series of rectifiers and typical use conditions of IF e 16A and VR e 200V and for two different junction temperatures of 25 C and 125 C Theory not only predicts but it has also been experimentally verified that these parameters are relatively independent of IF so only one value of the latter suffices Any three of the four Figures 10­13 completely specifies the reverse recovery behavior of the rectifier Since S and Trr vary the least over the plotting dIF dt range it is convenient to formulate reverse recovery energy loss P in microwatts in terms of the circuit parameters VR and dIF dt VR Pe where e minority carrier lifetime Wi e epitaxial thickness Da e ambipolar diffusion constant u The blocking voltage rating of the rectifier primarily determines Wi but for a given Wi note that a short minority lifetime not only decreases IR(REC) but happily increases S These two key parameters are plotted as a function of minority carrier lifetime in Figure 9 for dIF dt e 100 A ms and TJ e 25 C As has been noted before the minority carrier lifetime had been targetted for 20 ns­30 ns to minimize VF and this choice has resulted in a typical value of S e 0 65 and IR(REC) e 1 5A dt J
2S dIF f 1 SJ
Strr
a 2 10b3 (mW) e peak reverse voltage VR dIF dt e ramp rate (A ms) e operating frequency (kHz) f 4 TL G 10062 ­ 11 FIGURE 9 Theoretical Plots of IR(REC) and S vs Minority Carrier Lifetime TL G 10062 ­ 14 FIGURE 12 Reverse Recovery Time of the FRM FRP1600 Series Rectifier Diodes TL G 10062 ­ 12 TL G 10062 ­ 15 FIGURE 10 Reverse Recovery Current for the FRM FRP1620 Series Rectifiers FIGURE 13 Softness Factor S for the FRM FRP1600 Series Rectifier Diodes Example Calculate the reverse recovery power loss for the FRP1620 rectifier running at IF e 16A VR e 100V dIF dt e 100 A ms TJ e 125 C f e 75 kHz From Figures 12 and 13 trr e 56 ns and S e 0 29 Substituting these values in the above equation Pe (100V) (100 A ms)(75 kHz) (0 29)(56 ns) 2 b 10 3 mW (2)(0 29) 1 a 0 29 TL G 10062 ­ 13 FIGURE 11 Reverse Recovery Charge for the FRM FRP1600 Series Rectifier Diodes P e 0 205W There are may ways to shape the reverse recovery voltage spike The most simple and still most popular is the RC snubber circuit connected across the primary of the transformer in Figure 1b This serves the dual purpose of suppressing voltage ringing and EMI due to the switching action of both the transistor and rectifier William McMurray has shown how to design an RC snubber to minimize voltage transients and or dV dt ramps just due to the diode reverse recovery current (Reference 4) and also how to de- ( 5 Optimizing the Ultra-Fast POWERplanar Rectifier Diode for Switching Power Supplies sign snubbers to minimize transistor power dissipation (Reference 5) But to date because the RC snubber plays a major role in reducing EMI its design tends to be empirical rather than theoretical CONCLUSION This application note has pointed out the major considerations in designing an ultrafast reverse recovery rectifier and shown that the control of minority carrier lifetime is the key in arriving at an optimum device Because the diode contributes to EMI its reverse recovery behavior must be carefully controlled and characterized in order to guarantee similar performance from lot to lot REFERENCES 1 S K Ghandhi Semiconductor Power Devices (NYC) John Wiley and Sons p 110 ff 2 F Berz ``Ramp Recovery in p-i-n Diodes'' Solid-State Electronics Vol 23 pp 783 ­ 792 3 C M Hu Private Communication 4 W McMurray ``Optimum Snubbers for Power Semiconductors'' IEEE Trans on Industry Applications Vol 1A ­ 8 No 5 Sept Oct 1972 pp 593 ­ 600 5 W McMurray ``Selection of Snubbers and Clamps to Optimize the Design of Transistor Switching Converters'' PESC 1979 Conference Record pp 62 ­ 74 LIFE SUPPORT POLICY NATIONAL'S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF NATIONAL SEMICONDUCTOR CORPORATION As used herein 1 Life support devices or systems are devices or systems which (a) are intended for surgical implant into the body or (b) support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in a significant injury to the user 2 A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness AN-557 National Semiconductor Corporation 1111 West Bardin Road Arlington TX 76017 Tel 1(800) 272-9959 Fax 1(800) 737-7018 National Semiconductor Europe Fax (a49) 0-180-530 85 86 Email cnjwge tevm2 nsc com Deutsch Tel (a49) 0-180-530 85 85 English Tel (a49) 0-180-532 78 32 Fran ais Tel (a49) 0-180-532 93 58 Italiano Tel (a49) 0-180-534 16 80 National Semiconductor Hong Kong Ltd 13th Floor Straight Block Ocean Centre 5 Canton Rd Tsimshatsui Kowloon Hong Kong Tel (852) 2737-1600 Fax (852) 2736-9960 National Semiconductor Japan Ltd Tel 81-043-299-2309 Fax 81-043-299-2408 National does not assume any responsibility for use of any circuitry described no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications